Frequency Interleaved Directly Detected Optical OFDM for Next-Generation Optical Access Networks Lenin Mehedy, Masuduzzaman Bakaul, Ampalavanapillai Nirmalathas NICTA Victoria Research Laboratory Department of Electrical and Electronic Engineering, The University of Melbourne Victoria, 3010, Australia [email protected], [email protected], [email protected] Abstract— We theoretically analyze and demonstrate that spectral efficiency of a conventional direct detection based optical OFDM system (DDO-OFDM) can be improved significantly using frequency interleaving of adjacent DDOOFDM channels in future optical access systems.

I.

INTRODUCTION

Optical orthogonal frequency division multiplexing (OOFDM) brings the benefit of electronic equalization and robustness against multi-path fading of legacy wirelessOFDM systems into the optical domain to combat against fiber impairments, such as chromatic dispersion and polarization mode dispersion (PMD) and achieved impairments-tolerant ultra high speed optical systems [1]. Depending on the detection mechanisms, O-OFDM systems can be broadly categorized into two sub-groups namely, coherent O-OFDM (CO-OFDM) and directly detected (incoherent) O-OFDM (DDO-OFDM). Among them, COOFDM systems found to be more complex and expensive, as they require additional signal conditioning devices both in the transmitting and receiving ends [2]. On the other hand, a DDO-OFDM system offers simpler transmitter and receiver architectures [3]. Therefore it has the potential to be used in next-generation optical access networks, targeting 40 Gb/s and 100 Gb/s over un-amplified and dispersion uncompensated links for simplicity and low-cost [4]. Since a DDO-OFDM system directly detects the signal using a square law photodetector, it must have a spectral gap between the optical carrier and OFDM signal band to accommodate subcarrier-to-subcarrier beat products, which otherwise would contaminate the actual data. At optimum operating conditions, this required spectral gap needs to be either equal to or greater than the OFDM signal bandwidth [4]. Therefore, in a DDO-OFDM system, at least half of the signal bandwidth remains unused, reducing the effective optical spectral efficiency enormously. To utilize the unused spectral gap of a DDO-OFDM system a frequency interleaving method has been proposed, where two neighbouring DDOOFDM channels are overlapped in such a way that the mandatory spectral gap of a channel is being occupied by the OFDM signal band of its neighbour and thereby enhancing the effective spectral efficiency significantly [5, 6]. In this paper, we demonstrate such a frequency interleaved DDOOFDM system both by theoretical analysis and numerical

simulations. Our results show that, frequency interleaving of two adjacent DDO-OFDM systems each carrying 24 Gb/s pseudo-random-bit-sequence (PRBS) data with a OFDM signal band of 10 GHz (actual signal bandwidth is 25 GHz including the spectral gap), can increase the spectral efficiency up to 50% over a standard DDO-OFDM system. II.

SYSTEM DESCRIPTION

In this section we at first describe the effective spectral efficiency the proposed system followed by the theoretical analysis of such a system. A. Effective Spectral Efficiency of Interleaved Optical OFDM System As shown in Fig. 1, a conventional DDO-OFDM channel modulated in optical single sideband with carrier (OSSB+C) format with an effective bandwidth of B can be redesigned by frequency interleaving of two adjacent OSSB+C formatted DDO-OFDM channels where upper sideband (Band-1) of one channel falls within the mandatory spectral gap between the lower sideband (Band-2) and optical carrier of other channel and vice versa. In order to facilitate demultiplexing at the receiver, the OFDM signal band of these channels needs to be surrounded by spectral gaps as shown in Fig. 1. Therefore, each channel will have an OFDM signal bandwidth of kB rather than ½B, where ¼ < k < ½ such that total OFDM bandwidth is greater than half of the total system bandwidth. The effective spectral efficiency improvement is then given by (1).

1⎞ ⎛ γ = 2 ⎜ 2k − ⎟ × 100 % 2⎠ ⎝

Figure 1. Frequency interleaved DDO-OFDM system

(1)

Figure 2. Interleaved DDO-OFDM systems (a) schematic optical spectra , (b) schematic RF spectra after direct detection

Here γ is the normalized effective spectral efficiency improvement compared to a conventional system. Hence with k = 37.5%, the total OFDM bandwidth in the frequency interleaved system would be 75% of the available system bandwidth, which is a 50% increase compared to a conventional DDO-OFDM system.

of the local oscillator, peak amplitude of the electric field of the laser with a value of unity, optical carrier frequency, laser phase noise, optical modulation index, and modulator bias phase shift at the second O-OFDM transmitter. For simplicity of the following calculations, let us assume that A1_ carrier (t0 ) , A1_ data (t0 ) represent respectively the first

B.

and

Theory of Operation j ⎡ 2π f t + ϕ ( t ) ⎤

Modulating a laser E1e ⎣ 1 0 1 0 ⎦ with a radio frequency (RF) modulated OFDM signal [7], the OSSB+C formatted O-OFDM signal with only the upper sideband can be represented as

θ ⎡ ⎤ A1 (t0 ) = exp j ⎢ 2π f1t0 + 1 ( t0 ) + ϕ1 ( t0 ) ⎥ 2 ⎣ ⎦ ⎛ m + 1 ∑ ck ⎜⎜ exp 4 k =1 ⎜ ⎝ N SC

where,

θ1 ( t0 )

⎡ 2π ( f1 + f k + f RF 1 ) t0 ⎤⎞ ⎥⎟ j⎢ ⎢ +ϕ ( t ) + θ1 ( t ) + ϕ ( t ) ⎥ ⎟ 0 1 0 ⎢⎣ RF 1 0 ⎥⎦ ⎟⎠ 2

(2)

are respectively the total number of orthogonal

subcarriers in the system, the information symbol at k-th subcarrier, radio frequency of the local oscillator, phase noise of the local oscillator, peak amplitude of the electric field of the laser with a value of unity, optical carrier frequency, laser phase noise, optical modulation index, and modulator bias phase shift of the O-OFDM transmitter. Similarly, j ⎡ 2π f t + ϕ ( t ) ⎤

modulating another laser E2 e ⎣ 2 0 2 0 ⎦ and taking the lower sideband and the carrier, the output of the other DDOOFDM channel can be represented as,

where,

θ 2 ( t0 )

components of (2)

and

A2 _ carrier (t0 ) ,

A2 _ data (t0 ) represents respectively the first and second components of (3). After multiplexing these two DDOOFDM channels, the schematic spectrum of the frequency interleaved system looks like Fig. 2 (a). Now, for detecting the signal A1 (t0 ) , we assume that the signal A2 (t0 ) is attenuated by two cascaded ideal rectangular filter with stop band attenuation (SBA) factor of V, where V is the ratio of the attenuated signal power to the actual signal power. Then the output of the square law photodetector is given by,

v(t0 ) = R × [ A1 (t0 ) + VA2 (t0 ) ] × ⎡⎣ A*1 (t0 ) + VA*2 (t0 ) ⎤⎦

N SC , ck , f RF 1 , φRF 1 ( t0 ) , E1 , f1 , ϕ1 ( t0 ) , m1 and

θ ⎡ ⎤ A2 (t0 ) = exp j ⎢ 2π f 2t0 + 2 ( t0 ) + ϕ 2 ( t0 ) ⎥ 2 ⎣ ⎦ ⎛ ⎡ 2π ( f 2 − f k − f RF 2 ) t0 ⎤⎞ m N SC ⎜ ⎥⎟ + 2 ∑ d k ⎜ exp j ⎢ ⎢ −ϕ ( t ) + θ 2 ( t ) + ϕ ( t ) ⎥ ⎟ 4 k =1 ⎜ 0 2 0 ⎢⎣ RF 2 0 ⎥⎦ ⎟⎠ 2 ⎝

second

(3)

d k , f RF 2 , φRF 2 ( t0 ) , E2 , f 2 , ϕ 2 ( t0 ) , m2 and are respectively the information symbol at k-th

subcarrier, radio frequency of the local oscillator, phase noise

where R is the responsivity of the photodetector and * denotes complex conjugate. Assuming V to be very small and neglecting higher order V terms, the detected signal is given by (4) and shown schematically in Fig. 2 (b) for clarification. ⎡⎛ A1_ carrier (t0 ) A1_* carrier (t0 ) + A1_ carrier (t0 ) A1_* data (t0 ) ⎞ ⎤ ⎢⎜ ⎥ ⎟ * * ⎜ ⎟ ⎢⎝ + A1_ data (t0 ) A1_ carrier (t0 ) + A1_ data (t0 ) A1_ data (t0 ) ⎠ ⎥ ⎢ ⎥ * * ⎢ ⎛ A1_ carrier (t0 ) A2 _ carrier (t0 ) + A1_ carrier (t0 ) A2 _ data (t0 ) ⎞ ⎥ v ( t0 ) = R ⎢ ⎜ ⎟⎥ * * ⎢ ⎜ + A1_ data (t0 ) A2 _ carrier (t0 ) + A1_ data (t0 ) A2 _ data (t0 ) ⎟ ⎥ ⎟⎥ ⎢ +V ⎜ * * ⎢ ⎜ + A2 _ carrier (t0 ) A1_ carrier (t0 ) + A2 _ carrier (t0 ) A1_ data (t0 ) ⎟ ⎥ ⎟⎟ ⎥ ⎢ ⎜⎜ * * ⎢⎣ ⎝ + A2 _ data (t0 ) A1_ carrier (t0 ) + A2 _ data (t0 ) A1_ data (t0 ) ⎠ ⎥⎦

(4)

Now, considering only the desired detected signal in (4) and after some calculations we get (5), where the second component of (5) is the desired OFDM signal modulated on a RF and the first component is the complex conjugate of the second component or the image signal. As shown in Fig. 2 (b), some of the intermodulation products and OFDM signal band of the second channel falls within the desired signal bandwidth and corrupts the OFDM signal of the first channel unless the filters effectively suppress the second channel. Therefore, the effectiveness of proposed system largely depends on the filter’s stop band attenuation parameter (V) that enables successful suppression of the unwanted signal components.

Figure 3. Simulation setup of the interleaved DDO-OFDM system

v′(t0 ) = R ⎣⎡ A1 _ carrier (t0 ) A1 _ data (t0 ) + A1 _ data (t 0 ) A1 _ carrier (t0 ) ⎦⎤ *

*

⎡ m1 N ⎤ ⎢ 4 ∑ ck ( exp − j [ 2π ( f k + f RF 1 ) t0 + ϕ RF 1 ( t0 ) ]) ⎥ k =1 ⎥ = R⎢ ⎢ m1 N ⎥ ⎢ + 4 ∑ ck ( exp j [ 2π ( f k + f RF 1 ) t0 + ϕ RF 1 ( t0 ) ]) ⎥ ⎣ ⎦ k =1 SC

(5)

SC

III.

SIMULATION SETUP

Given a 25 GHz channel grid, we have chosen two channels each with an OFDM bandwidth of 9.5 GHz. After frequency interleaving, a total OFDM signal bandwidth of 19 GHz is achieved, which is 50% higher compared to a conventional system having a maximum OFDM bandwidth of 12.5 GHz. Functionality of the proposed system is then verified with a simulation model using VPITransmissionMakerTM7.6 as shown in Fig. 3, where each of the single channel DDO-OFDM system comprises of a zero filled centre subcarrier, surrounded by 194 orthogonal subcarriers. Among these subcarriers, 186 subcarriers carry 8quadrature amplitude modulation (8-QAM) encoded PRBS data and 8 (eight) equally spaced pilot subcarriers carry binary phase shift keyed (BPSK) encoded pilot symbols. Then the OFDM signal with necessary oversampling is generated by a 256 point inverse fast Fourier transform (IFFT) module by filling the remaining subcarriers with zeros. The last 32 samples of each OFDM symbol (12.5% of the IFFT size) are added in the beginning of the symbol as a cyclic prefix (CP). The generated block of complex valued OFDM symbols are then serialized, separated into in-phase (I) and quadrature (Q) components, and converted to analog wave forms using two Digital-to-Analog (DAC) modules with a sampling rate of 12.5 GS/s such that after upconverting these I and Q signals to an intermediate RF frequency (18.75 GHz), an OFDM signal with a bandwidth of 9.5 GHz is generated that carries 24 Gb/s of data. This RF

Figure 4. (a) Optimization of filter F1, (b) SNR of different subcarriers with different SBA of filter F1

modulated OFDM signal is then modulated over an optical carrier using a Mach-Zehnder modulator (MZM) having insertion loss of 4 dB and bias voltage close to its transmission null [3]. The first O-OFDM transmitter generates an optical carrier using a laser module having 5 MHz laser linewidth, 3 mW average power and emission frequency of 193.1 THz whereas the second transmitter uses a similar laser module but having an emission frequency set at 25 GHz apart of the first one. Lower side band of the first channel and the upper side of the second channel are then removed using suitably tuned two 50 GHz optical band pass filters (OBPF). Two such 24 Gb/s OSSB+C formatted OOFDM signals are then combined using an optical power combiner to generate the desired interleaved DDO-OFDM signal and amplified to an optical power of 0 dBm. The composite signal is then transmitted over 25 km single mode fiber (SMF) with dispersion of 17 ps/nm/km, attenuation of −20 0.2 dB/km, nonlinear index of 2.6 × 10 m2/W, and PMD of 0.1 ps/√km. At the receiver side one of these two signals (i.e. Band-1) is demultiplexed using two cascaded 12.5 GHz optical band stop filters (OBSF) that suppress the unwanted data band and optical carrier. The DDO-OFDM receiver then directly detects the received signal using a PIN photodetector with a 70% responsivity factor and down-converts the OFDM signal to baseband using a RF I-Q down-converter. The baseband I, Q signals are then digitized using two analog-todigital-converters (ADCs) and passed to the electrical OFDM receiver to recover transmitted data bits. The electrical OFDM receiver performs the necessary digital signal processing including cyclic prefix removal, FFT processing, channel equalization and phase noise compensation [2], [7]. A total of 200 OFDM symbols are transmitted in the simulation among which first two symbols are used for channel estimation and zero-forcing equalization.

IV.

SIMULATION RESULTS

Performance of the interleaved DDO-OFDM system is investigated by calculating the error vector magnitude (EVM) of the data band (Band-1) of the first DDO-OFDM channel (channel-1). At first, performance is measured by varying the stop band attenuation (SBA) of the filter (F1) that suppresses the unwanted data band (Band-2) of the second channel (channel-2) with the filter F2 disabled, as shown in Fig. 3. Simulation results as shown in Fig. 4 (a) confirm that SBA of 30 dB for F1 is sufficient. SNR of the data subcarriers are shown in Fig. 4 (b) with respect to that of a single channel DDO-OFDM system and varying SBA factors. Fig. 4b

Figure 5. (a) Optimization of filter F2, (b) SNR of different subcarriers with different SBA factor of F2.

validates the theoretical prediction that with lower SBA factor, the data subcarriers near dc are more affected than the subcarriers at higher frequencies when the unwanted data band is not sufficiently suppressed. Fig. 4b also shows that overall SNR of the data subcarriers are below that of the single channel 24 Gb/s DDO-OFDM system. This is because the unwanted optical carrier of the channel-2 has not been suppressed yet. Therefore, setting the SBA of filter F1 at the 30 dB, the SBA of the second filter F2 is varied that suppresses the unwanted optical carrier of channel-2 and performance results are shown in Fig. 5 (a) and 5 (b). Fig. 5 (b) shows that with higher SBA of filter F2, the overall SNR of the data subcarriers improve as expected but the higher frequency subcarriers are still indicating low SNR. The reason is that the laser phase noise of the channel-2 lowers the SNR of the higher frequency data subcarriers of channel-1 as shown in Fig. 6 (a) and 6 (b) respectively for laser line-widths (LW) of 5 MHz and 100 kHz. Therefore SNR of the higher frequency subcarriers of channel-1 can be improved and matched with the benchmarked single channel system either by suppressing the laser phase noise of channel-2 using another filter T1 at the transmitter side before interleaving, as shown in Fig. 6 (c) or by using narrow linewidth lasers, as shown in Fig. 6 (d). Fig. 7 shows the back-to-back performances with commercially available non-rectangular type 12.5 GHz filters having a 10 GHz bandwidth at the 30 dB attenuation point. Fig. 7 confirms that there is a maximum penalty of only 0.3 dB for non-rectangular type filters. Transmission performance of the system after 25 km SMF without any inline amplification and dispersion compensation is shown in Fig. 8. Fig. 8 shows that system performance is similar with and without interleaving. It also shows that the receiver sensitivity is -27.5 dBm after 25 km SMF that

Figure 7. Performance with different types of optical filters

achieves a power margin of 16.5 dB, which is enough for 1: 32 split passive optical networks. V.

Theoretical analysis and simulation results suggest that frequency interleaving of two neighbouring DDO-OFDM channels can improve the spectral efficiency of a conventional DDO-OFDM system by 50%. After successful transmission over 25 km of un-amplified and dispersion uncompensated link, such a frequency interleaved DDOOFDM system with a bit rate of 48 Gb/s within 25 GHz bandwidth can achieve a power margin of 16.5 dB, implying its potential for future high speed optical access networks. REFERENCES [1]

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Figure 6. (a) and (b) show optical spectrum with laser linewidth of 5 MHz and 100 kHz respectively; (c) and (d) show SNR of subcarriers of channel-1 when laser phase noise is reduced with another filter T1 at the transmitter and when 100 kHz lasers are used

CONCLUSION

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Figure 8. Transmission after 25 km SMF without inline amplification