pole antenna for wideband and multiband operation of wireless communications, IEE Proc Microwaves Antennas Propagat 151 (2004), 473– 476. 5. C.M. Su, K.L. Wong, W.S. Chen, and Y.T. Cheng, A microstripcoupled printed inverted-F monopole antenna, Microwave Opt Technol Lett 43 (2004), 470 – 472. 6. E. Lee, P.S. Hall, and P. Gardner, Compact wideband planar monopole antenna, Electron Lett 35 (1999), 2157–2158. © 2005 Wiley Periodicals, Inc.

ELECTROMAGNETIC BANDGAP ENHANCEMENT USING THE HIGHIMPEDANCE PROPERTY OF OFFSET FINITE-GROUND MICROSTRIP LINE Sheng Sun and Lei Zhu School of Electrical and Electronic Engineering Nanyang Technological University Singapore 639798 Received 15 June 2005

Figure 8 Measured peak gain of the proposed antenna: (a) 2450-MHz band (2400 –2484 MHz); (b) 5800-MHz band (5725–5875 MHz)

hand, the gain variation for the 5.25-GHz band is about 3.21– 4.3 dBi and for the 5.8-GHz band is about 3.68 –5 dBi. For the dual-band antenna design aspect, the lower resonant mode has impedance bandwidth (2:1 VSWR) of 195 MHz (2370 – 2565 MHz), which covers the 2.45-GHz band for WLAN operation (2400 –2484 MHz). The higher bandwidth covers ISM operation (5725–5875 MHz), with 189 MHz (5679 –5808 MHz). They demonstrate the radiation patterns of the proposed dual-band antenna for 2.45- and 5.8-GHz operations, respectively. The measured results indicate that the proposed antenna exhibits nearly omnidirectional radiation patterns at the x–y plane for every single operating band. Generally, the peak gain of the proposed dualband antenna exhibits fairly stable antenna gain for applications. REFERENCES 1. M. Ali and G.J. Hayes, Small printed integrated inverted-F antenna for Bluetooth application, Microwave Opt Technol Lett 33 (2002), 347– 349. 2. S.H. Yeh and K.L. Wong, Integrated F-shaped monopole antenna for 2.4/5.2-GHz dual-band operations, Microwave Opt Technol Lett 34 (2002), 24 –26. 3. Y.F. Lin, H.D. Chen, and H.M. Chen, A dual-band printed L-shaped monopole for WLAN applications, Microwave Opt Technol Lett 37 (2003), 214 –216. 4. W.C. Liu, W.R. Chen, and C.M. Wu, Printed double S-shaped mono-

ABSTRACT: The high-impedance property of an offset finite-ground microstrip line is utilized to construct an improved transmission-line electromagnetic bandgap (EBG) structure with enhanced bandgap width and attenuation. Using the self-calibrated method of moments (MoM), the two effective per-unit-length parameters are firstly extracted to demonstrate the fundamental frequency-dispersive characteristics of the guided wave in this EBG. Then, the transmission coefficients of the two finite-length EBG circuits with ideal impedance matching at two ports are characterized, thus exhibiting the distinctive bandstop and bandpass behaviors as the guided-wave propagates across this EBG with finiteextended region. Finally, a five-cell EBG circuit, fed with the standard 50⍀ feed lines, is further modeled to give an evident verification of its enhanced EBG performance. © 2005 Wiley Periodicals, Inc. Microwave Opt Technol Lett 47: 543–546, 2005; Published online in Wiley InterScience (www. interscience.wiley.com). DOI 10.1002/mop.21225 Key words: periodic structure; electromagnetic bandgap; guided-wave; offset finite-ground microstrip line; per-unit-length transmission parameter 1. INTRODUCTION

During the past decades, periodic structures have been generating continuous interest for the development of RF and microwave integrated circuits because of their slow-wave and frequency filtering properties. With series-inductive or shunt-capacitive loading in periodic intervals, nonuniform planar transmission lines exhibit stopband characteristics in a certain range [1]. This periodic structure with stopband was recently referred to as the photonic bandgap (PBG) [2] or electromagnetic bandgap (EBG) [3]. In past years, various EBG structures have been developed for exploring advanced circuit blocks [2], suppressing unwanted modes in shielded planar circuits [3], eliminating harmful noises in highspeed printed circuit boards [4], and so on. On the other hand, planar EBG structures have been more comprehensively investigated in terms of the two effective perunit-length transmission parameters of periodic structures with infinite extension [5–7]. In particular, nonzero attenuation constant and imaginary characteristic impedance in the stopband were reported in [7] in order to display the complete guided-wave characteristics of microstrip EBG. In this paper, our effort is focused on achieving enhanced EBG performance using the high-imped-

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of the two-port ABCD matrix defined at the interfaces between the periodic and uniform FGMSL [7]: Z0 ⫽

冑

B C

and

cosh共␥L兲 ⫽

A⫹D . 2

(1)

3. RESULTS AND DISCUSSION

Figure 1 The 3D geometry of periodic electromagnetic bandgap (EBG) finite-ground microstrip line: (a) unit-cell; (b) N-cell

ance property of offset finite-ground microstrip line (FGMSL). Figure 1 depicts the 3D geometry of the proposed EBG structure based on the periodic FGMSL. Using the modeling technique in [8], the effective propagation constant and characteristic impedance of this EBG structure are extracted in order to demonstrate the enhanced guided-wave bandgap behavior with widened stopband. The magnitude and phase quantities of the transmission coefficients for a five-cell EBG circuit are then displayed under ideal impedance matching at the two ports. This EBG circuit with two 50⍀ feed lines is further constructed and characterized in order to demonstrate the enhanced EBG performance existed in an actual planar EBG circuit.

Figures 2(a) and 2(b) depict the two extracted complex transmission parameters of the periodic FGMSL with different strip widths of W 1 ⫽ 0.6 and 1.2 mm, respectively. In these two cases, two narrow strip conductors of the offset FGMSL are always put at the extremely outer edges of the wide strip conductors, as shown in Figures 1(a) and 1(b). In the low-frequency range, the normalized propagation constant ( ␥ /k 0 ) has the only imaginary part and this normalized phase constant ( ␤ /k 0 ) is significantly raised as W 1 is widened from 0.6 to 1.2 mm. This is because the widened W 1 leads to enlargement of the coupling between the lower and upper strip conductors in the symmetrical FGMSL, while the increased offset distance (W 1 ⫺ W 2 ) tremendously reduces this coupling in the offset FGMSL. As the frequency increases, the guided wave becomes slow in the propagation region and then moves itself into the attenuation region, namely, the bandgap in [2– 4], with the nonzero imaginary impedance and attenuation constant, as also described in [7, 9]. Looking at the two sets of graphs together, we can observe that the bandgap has really gained a large enhancement from 41% to 81% in fractional bandwidth. Beyond this

2. EBG GEOMETRY AND PARAMETRIC EXTRACTION

As illustrated in Figure 1, the proposed EBG structure is constructed on the basis of the FGMSL [8] with periodic configuration on the upper and lower finite-width strip conductors. Looking at the schematic of a single EBG unit cell in Figure 1(a), the two lower and upper strip conductors in the central part are simultaneously narrowed and then transversely separated so as to construct an offset FGMSL with high characteristic impedance. Meanwhile, the two strips at the two sides are equally widened to form an FGMSL with low characteristic impedance. In this way, the former one can be perceived as an equivalent series-inductive element, while the latter as equivalent to a shunt-capacitive element. It can be intuitively understood that the equivalent seriesinductive quantity in the offset FGMSL can be largely enhanced by separating the two strip conductors away from each other. In the modeling, a periodic FGMSL with cell number N is linked with the two uniform FGMSL feed lines and its performance can be in general characterized using the self-calibrated method of moments (MoM) [8]. As pointed out in [7], this N-cell periodic FGMSL or EBG structure can be considered as an equivalent uniform transmission-line section with length L ⫽ NT, which may have frequency-dispersive, full-attenuation or fulltransmission performances. To investigate the fundamental guided-wave characteristics in this infinite-extended EBG, the two effective per-unit-length parameters of this artificial transmission line, that is, complex propagation constant ( ␥ ⫽ ␣ ⫹ j ␤ ) and complex characteristic impedance (Z 0 ⫽ Re(Z 0 ) ⫹ j Im(Z 0 )), need to be derived as discussed in [5–7]. They can be explicitly expressed in terms of the four calculated elements, A, B, C, and D,

544

Figure 2 Extracted per-unit-length complex transmission parameters of infinite-length periodic FGMSL: (a) normalized propagation constants; (b) characteristic impedances

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this stopband. We can see here that the bandgap in W 2 ⫽ 1.2 mm is much wider in bandwidth and deeper in attenuation than that in W 1 ⫽ 0.6 mm. This phenomenon is fully controlled by the frequency region and magnitude of nonzero ␣, as plotted in Figure 2(a). Subsequently, it will be exhibited that the guided-wave bandgap characteristics with nonzero ␣ actually determine the stopband width and insertion loss of the finite-length EBG circuit with 50⍀ feed lines. The phase ⌽21 of these two EBG structures is plotted in Figure 3(b). Regardless, this parameter with negative quantity decreases in the lowpass band, becomes the constant of ⫺5␲ in the whole bandgap, and then decreases again in the upper passband. Finally, the above five-cell periodic EBG structures are connected with the actual 50⍀ feed line at the two truncated terminals. Similarly, the relevant scattering parameters can be calculated in terms of the 50⍀ source/load impedance, the above-extracted Z 0 and ␥ as well as the finite-length L. Figures 4(a) and 4(b) show the calculated insertion loss 兩S 21 兩 and return loss 兩S 11 兩 based on the simple transmission-line theorem together with those obtained by directly simulating the whole layouts of these EBG circuits using Agilent’s Momentum. Looking at Figures 3(a) and 4(a), the bandstop or bandgap in both cases appears in almost the same frequency region. Several ripples observed in the low and high passbands in Figure 4(a) are mainly contributed by the multiple reflections between the two interfaces along the truncated EBG structure. But the results in Figure 4 show us again that the EBG

Figure 3 Magnitude and phase of the transmission coefficients of fivecell periodic FGMSL EBG structures with perfect impedance matching at two terminals: (a) magnitude (兩S 21 兩); (b) phase (⌽21)

bandgap, the guided wave enters into its next passband region [7, 9]. Let us now consider the guided wave propagating across the truncated finite-cell periodic FGMSL that is assumed to be ideally matched with the source and load impedances. This arrangement is made to illustrate firstly the propagation performance in this EBG structure without receiving any unexpected mismatching effects at two interfaces. On the basis of the cascaded transmission-line theorem, the transmission coefficient (S 21 ) of this periodic EBG transmission line with length L ⫽ NT can be simply expressed as follows, where the impedance (Z 0 ) disappears due to ideal matching: S 21 ⫽ e ⫺␣L⫺j␤L.

(2)

Furthermore, the magnitude 兩S 21 兩 and phase ⌽21 can be separately derived in an explicit format, given by 兩S 21兩 ⫽ e ⫺␣NT

and ⌽21 ⫽ ⫺␤NT.

(3)

Figures 3(a) and 3(b) show the two sets of these parameters for the above-discussed periodic EBG structures with five cells, that is, N ⫽ 5. In the low and high passbands, the magnitude 兩S 21 兩 is observed in Figure 3(a) as the unity constant in both cases, since the attenuation constant ␣ is consistently equal to zero. In the bandgap region between them, this parameter decreases and approaches the minimum as the frequency is moved to the center of

Figure 4 Scattering parameters of five-cell periodic FGMSL EBG circuits with 50⍀ external feed lines: (a) transmission coefficient (兩S 21 兩); (b) reflection coefficient (兩S 11 兩)

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performance has been really enhanced, as W 1 is widened from 0.6 to 1.2 mm. 4. CONCLUSION

In this work, a novel periodic electromagnetic bandgap structure has been presented, characterized, and implemented. Using the high-impedance property of the offset finite-ground microstrip line, the bandwidth and attenuation depth of the concerned guidedwave bandgap have been quantitatively confirmed to gain the significant enhancement. The two effective per-unit-length parameters were derived in order to display the fundamental guidedwave characteristics of the FGMSL EBG with infinite-extended length [5–7]. The scattering parameters of the finite-cell EBG circuits with varied feed-line impedances were further investigated and found to exhibit bandstop behaviors as a two-port filtering circuit [2– 4]. REFERENCES nd

1. R.E. Collin, Foundations for microwave engineering, 2 ed., McGrawHill, New York, 1992. 2. F. Yang, R. Coccioli, Y. Qian, and T. Itoh, Planar PBG structures: Basic properties and applications, IEICE Trans Electron E83-C (2000), 687– 696. 3. D. Dawn, Y. Ohashi, and T. Shimura, A novel electromagnetic bandgap metal plate for parallel plate mode suppression in shielded structures, IEEE Microwave Wireless Compon Lett 12 (2002), 166 –168. 4. R. Abhari and G.V. Eleftheriades, Metallo-dielectric electromagnetic bandgap structures for suppression and isolation of the parallel-plate noise in high-speed circuits, IEEE Trans Microwave Theory Tech 51 (2003), 1629 –1639. 5. C.-K. Wu, H.-S. Wu, and C.-K.C. Tzuang, Electric-magnetic-electric slow-wave microstrip line and bandpass filter of compressed size, IEEE Trans Microwave Theory Tech 50 (2002), 1996 –2004. 6. S.-G. Mao and M.-Y. Chen, Propagation characteristics of finite-width conductor-backed coplanar waveguides with periodic electromagnetic bandgap cells, IEEE Trans Microwave Theory Tech 50 (2002), 2624 – 2628. 7. L. Zhu, Guided-wave characteristics of periodic microstrip lines with inductive loading: slow-wave and bandstop behaviors, Microwave Opt Technol Lett (2004), 77–79. 8. S. Sun and L. Zhu, Unified equivalent circuit model of finite-ground microstrip line open-end discontinuities using MoM-SOC technique, IEICE Trans Electron E87-C (2004), 828 – 831. 9. L. Zhu, Guided-wave characteristics of periodic coplanar waveguides with inductive loading– unit-length transmission parameters, IEEE Trans Microwave Theory Techniques 51 (2003), 2133–2138. © 2005 Wiley Periodicals, Inc.

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ANTENNA-COUPLED MICROBOLOMETERS ON A SILICONNITRIDE MEMBRANE F. J. Gonza´lez,1 B. Ilic,2 and G. D. Boreman3 Instituto de Investigacio´n en Comunicacio´n Optica Universidad Auto´noma de San Luis Potosı´ Alvaro Obrego´n 64 San Luis Potosı´, SLP, Me´xico 2 CNF, Cornell University Ithaca, NY 3 UCF, College of Optics & Photonics Orlando, FL

1

Received 14 June 2005

ABSTRACT: Square-spiral antenna-coupled microbolometers made of gold and chrome are fabricated on a silicon-nitride membrane. The membrane gives mechanical stability and thermal isolation to the detectors so that they can be used as picture elements in commercial infrared-imaging systems. The fabricated devices are measured under room pressure and under vacuum, giving D* values of 3 ⫻ 106 cm公Hz/W and 1.7 ⫻ 107 cm公Hz/W, respectively. By using vanadium-oxide, values well above 1 ⫻ 108 cm公Hz/W for D* can be obtained. Variations to the fabrication process that will allow the use of vanadium oxide are suggested. © 2005 Wiley Periodicals, Inc. Microwave Opt Technol Lett 47: 246 –248, 2005; Published online in Wiley InterScience (www. interscience.wiley.com). DOI 10.1002/mop.21226 Key words: antenna-coupled detectors; microbolometers; silicon nitride membrane; responsivity; thermal impedance 1. INTRODUCTION

Antenna-coupled microbolometers have shown response times as low as 130 ns [1] and they can be used as picture elements in infrared-imaging systems [2, 3], which makes them a promising option for fast-frame-rate infrared-imaging applications. The main problem with antenna-coupled microbolometers is that their responsivity is lower than that required for commercial infraredimaging systems [4]. The responsivity of a microbolometer can be increased by using a bolometric material with a higher temperature coefficient of resistance (TCR) [5] or by increasing the thermal impedance of the device [4]. The highest thermal impedance will occur when the detector is completely isolated from the environment; this will reduce to zero the thermal conductivity of a device. Suspending a device on air above its substrate will increase its thermal impedance by eliminating heat conduction through the substrate [6] (Fig. 1); the standard procedure to make a suspended device is to fabricate it on top of a “sacrificial layer” which will later be selectively etched away, leaving just the patterned structure suspended on air. Silicon dioxide (SiO2) is widely used as a sacrificial layer because it is easy to deposit and can be etched away with hydrofluoric acid (HF), which will not etch silicon and therefore can be used as an etch-stop. Thin-film stress and stiction are the main problems associated with fabricating standing structures. Thin-film stress will make the suspended structures buckle and stiction will make the structures collapse due to surface tension of liquids during evaporation; this can be solved by using alternative rinsing procedures such as critical point drying or freeze drying [7]. These problems become important if we increase the area of the suspended devices, which occurs in infrared imaging appli-

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